反激的典型波形讲义

反激的典型波形反激变换器的例子Analysisof basicwaveforms基本波形分析The analysis of the basic waveformswill be done on a simula ted exampleof a flyback converteroperating in discontinuouscond uction mode.Typical drain-sourcevoltage waveform of the primary side switch is shown in Fig. 16.在电感电流断续模式下运行的反激变换器的典型一次侧 漏源极开关电压波形见图1 6 oFig. 16 Typical drain-sourcevoltage of the MOSFET in a flyback 图1 6反激变换器的典型漏源极电压Thesedrain-sourcevoltage waveformscan be theoretically distin guishedinto typical elementsDifferent physical phenomenanfluence the waveform at given time interval. Fig. 17 and Tab. 4 dem nstrate the main elementsof the voltage waveform. The superpos ion of all theseelementsresults in a typical drain-sourcevoltageshown in Fig. 16.这些漏源极电压波形能用典型的理论来描述。各个时间段有不同物理现象影响这些波形。图1 7和平台4描述了电压波形的主要原理。把这些原理按时序整合呈现由图1 6所示的典型漏源极电压。Fig. 17 Main elementsof the drain-sourcevoltage图1 7漏源极电压的主要原理Element 1:voltage fall during turning onElement 2:parasitic oscillaiion during turning on due to current spikeo 5 io-8原理2:在开通期间因寄生震荡产生的电流尖刺Element 3:voltage rise during turning off原理3 :关断期间的电压上升Elerr clam snubtoo制40M0IEP fc1一mt 4:ing voltage of*eri 1 m v nt n n m rir 导1.| t .1.1*1* 1i)* 1 .t t ii i1 ii i 1 ,， 56-10 S3 10原理4:缓冲电路的钳位电压Element 5:parasitic oscillation after clamping involving mainly the output capacitance of the MOSFET and the leakage inductance of the transformertoo ri1SO -109L1162 W*原理5 :钳位过程结束后主要由场效应晶体管输由电容和变压器漏感引起的寄生振荡Element 6:parasitic oscillation after flyback phase involving mainly the output capacitance of the MOSFET and the magnetization inductance of the transformer100原理6 :soo-so2 8-100产 3 2-10*-100磁芯存储磁能释放完毕后主要由场效应晶体管输由电容和变压器电感引起的寄生振荡Element 7: reflected voltage durirg he fl/tiack phe？eELsmerit B: main rectangular atgnal with Bia amplilud原理7:反激变换器释放磁能期间的反射电压原理8:与直流母线电压等幅的主要方波Tab. 4 Main elementsof the drain-sourcevoltage平台4漏源极电压的主要原理The spectrumof the whole drain-sourcewaveform (Fig. 16) is presentedn Fig. 18.图1 6所示的漏源极电压呈现的电磁干扰频谱见图1 8 ofrc+P .4Eton图2 0漏源极电压主要原理产生的电磁干扰频谱This method allows associatingcertain parts of the spectrum with their root causesj.e. the peak at 20 MHz in the spectrum of the drain-sourcevoltage is causedby the parasitic oscillation due to the output capacitanceof the MOSFET and the leakagei nductanceof the transformer.这种方法可以确定电磁干扰频谱中某些频点的来源，也就是说漏源极电压产生的电磁干扰频谱中的2。兆赫兹峰点是钳位过程结束后主要由场效应晶体管输由电容和变压器 漏感引起的寄生振荡产生的。The analysis of the drain current of the primary switch will b e done in the sameway. Fig. 21 demonstrates typical drain cur rent in a DCM flyback.对一次侧开关的漏极电流进行分析采用相同的方法。图21展示由一个工作于电感电流断续模式反激变换器的典型漏极电流。Fig. 21 Typical drain current in a flyback4iirtune, si*1tvr图2 1反激变换器的典型漏极电流This waveform can be presentedas a superpositionof the follo wing elements(Fig. 22 and Tab. 5). The superpositionof all thes e elementsresults in a typical drain current shown in Fig. 21.这个波形可以被看作是下列原理的叠加(图2 2和平台5) o全部这些波形的叠加整合结果变成图2 1所示的典型 漏极电流。s mT1 SpIf Np Itnif ，TUKpI upFig. 22 Main elementsof the drain current图2 2漏极电流的主要原理Element 1:苗 tndnqlE erf 曲阜 dsin cuirrnt原理1:漏极电流的主要三角波形Element 2;current spike during turning on due to parasitic capacitances of the circuit原理2 :在开关开通期间因寄生分布电容引起的电流尖刺原理3 :Element 3;parasiiic oscillation after clamping involving mainly the output capacitance of the MOSFET and the leakage inductance of the transformer钳位过程结束后主要由场效应晶体管输由电容和变压器漏感引起的寄生振荡0.040 027 02-0.04-0.06Element 4:parasitic oscillation after flyback phase involving mainly the output capacitance of the MOSFET and the magnetization inductance of 1he transformer0 06原理4:磁芯存储磁能释放完毕后主要由场效应晶体管输由电容和变压器电感引起的寄生振荡Tab. 5 Main elementsof the drain current平台5漏极电流的主要原理The spectrumof the whole drain current waveform (Fig. 21) is presentedin Fig. 23.全部漏极电流波形产生的电磁干扰频谱（图2 1 ）呈现在z freQiwflcy. Hi iTwFig. 23 Spectrumof the drain current (as shown in Fig. 22) 图2 3漏极电流产生的电磁干扰频谱(与图2 2相同)The spectraof the main elementsof the drain current can be found in Fig. 25. Fig. 24 is exactly the sameas Fig. 22 and ha s beenrepeatedfor better understanding.漏极电流主要原理产生的电磁干扰频谱见图2 5。图2 4和图2 2相同。5mTitrit HtiK, s j-kTritrFig. 24 Main elementsof the drain current= llaunu图2 4漏极电流的主要原理u/2 frequency, llz tlru?Fig. 25 Spectraof the main elementsof the drain current 图2 5漏极电流主要原理产生的电磁干扰频谱vs中 p -Sas 30As in case of drain-sourcevoltage this method allows to associ ate the elementsof the drain current waveform with its contribut ion to the whole spectrum.For example,the peak at 20 MHz inthe spectrumis causedby the parasitic oscillation due to the ou tput capacitanceof the MOSFET and the leakageinductanceof t he transformer.就象漏源极电压的例子那样，用这种方法也可以找由漏极电流的哪一部分对电磁干扰频谱产生影响。举例说明，2 0 兆赫兹的峰点是钳位过程结束后主要由场效应晶体管输由 电容和变压器漏感引起的寄生振荡产生的。This method of separatingthe waveform in time domain into i ts main elementshelps to find out what part of the spectrumin frequencydomain causedby what related physical phenomenaTh e separationinto main elementsshould be done in respectof reas onableeventsin the power circuit like on and off slopesoscillati ons, clamping,snubbering,reflectedvoltage, etc.这种在时域里对主要原理进行拆分的方法有助于我生产 生电磁干扰频段的干扰源。这种离析主要原理的手法有助于 合理审视电源电路里诸如变化速率、振荡、钳位、缓冲、反 射电压等过程。In this flyback exampleonly the primary switch has beenanal yzed as active sourceof electrical noise.Thereare also others, like secondaryside diodesor synchronousectifier, control IC (especial ly its gate drive), etc. In order to obtain more completeanalysis all theseinterferencesourceshave to be analyzed.在这个反激变换器里只对一次侧开关进行电磁噪声产生 的分析。但是还有其他的部分，象二次侧的二极管或同步整 流器、控制集成电路（尤其是它们的栅极驱动）等等。按顺 序分析将获得更完善的关于这些电磁干扰源的解析。However, it is impossibleto predict the conductedEMI spectr um using this approachdue to the fact, that only interferences。 urcesare consideredThereis no analysis of the spreadingpaths o f the interferencein this method.然而，这种方法不可能预知用频谱反映的电磁干扰的实际 行为，仅仅是干扰源被重视起来。在那里没有对分布参数产 生的干扰进行分析的方法。Neverthelessthe associationof harmonicsroot causewith the respectecphysical phenomenavill reducethe efforts of EMI reduc tion. The impact of the identified root causecan be reducednot only by filtering, but also by meansof influencing the root cause itself.不过，重视物理现象并不能成就电磁干扰的降低。降低干扰并不仅仅是滤波，也同样意味着干扰源自身的影响。Operationmodesof discontinuouslyback converter电感电流断续工作反激式变换器的运行模式The flyback converterrunning in discontinuousconductionmode can be operatedin hard switching or quasi resonant(or valley switching, or ZVS) mode regardingthe primary side switch. The differencebetweena hard switching and quasi resonantflyback co nverter is the turn on time point of the primary switch. In a ha rd switching modethe turning on of the MOSFET is not synchro nized with the drain-sourcevoltage value. This type of converters runs mainly in fixed frequencymode.电感电流断续工作的反激式变换器一次侧开关可工作于 硬开关或准谐振(或谷值开关或零电压开关)模式。硬开关 和准谐振反激变换器之间的差异在于一次侧开关的开启时 间点。在硬开关里场效应晶体管的开启波形拐点并不和漏源 极电压值同步。这种变换器大体上运行于固定频率模式。In a quasi resonantmodethe resonantcircuit determinedby t he output capacity of the MOSFET and the inductanceof the tr ansformerwill be utilized to switch on at lowest possiblevalue o f the drain-sourcevoltage. This circuit starts to oscillate at the e nd of the current flow through the secondaryside of the transfor mer, henceat the end of the flyback phase.The MOSFET will b e turned on at the minimum of this oscillation. The quasi resonan t approachuses this oscillation to achieveminimum voltage switch ing during turn on for the MOSFET. This operation mode runs a t a variable frequency.在准谐振模式里，由变压器电感和场效应晶体管输由电容 引起的谐振促使开关的开通时刻发生在漏源极电压的最小 值上。这种电路在电流从变压器二次侧流尽以后（反激回扫 过程结束）开始振荡。场效应晶体管将在振荡幅值的最小值 开启（谷值开通）。这种运行模式工作在可变的频率上。Higher amplitude of the oscillation results in lower drain sourc e voltage level at which the MOSFET turns on correspondingly wer switching lossesand higher efficiency of the system.更高幅值的振荡导致场效应晶体管更低的漏源极开通电 压幅值来产生更低的开关损耗和更高的系统效率。To achievehigh oscillation peaks,the designof the transformer has to be set to high reflectedvoltage. This increaseof the refle cted voltage results in a higher drain-sourcevoltage blocking MOS FET and longer duty cycles.要达到比较高的振荡电压峰值，变压器的反射电压必须设置的比较高。增加的反射电压导致使用更高漏源极击穿电压 的场效应晶体管和更大的开关占空比。Comparisonof three different flyback solutions has beenmade. All of them have beenoperation at 300 kHz, bus voltage of 40 0 V, output power of 120 W, output voltage of 16 V. Thesedesign included different modesof operationand different values of reflectedvoltage, resulting in different MOSFET s voltage ratings:比较现有的三种反激变换器。它们都工作在3 0。千赫兹，直流母线电压4 0 0伏特，输由功率1 2 0瓦特，输由电压1 6伏特。这些设计包含不同的运行模式和反射电压等级，因此使用不同电压等级的场效应晶体管： IHard switching flyback with CoolMOS 600V, reflectedvoltage of 100V硬开关反激变换器使用6 0 0伏特 CoolMOS 1 0 0伏 特反射电压Quasi resonantflyback with CoolMOS 600V, reflectedvoltage of 100V准谐振反激变换器使用6 0 0伏特 CoolMOS 1 0 0伏 特反射电压Quasi resonantflyback with CoolMOS 800V, reflectedvoltage of 390V准谐振反激变换器使用80。伏特CoolMOS 390伏特反射电压The clampingsnubbercircuit was set to the rated breakdownvoltage of the MOSFET (600 V and 800 V respectively).钳位缓冲电路被设定在场效应晶体管的额定击穿电压上（分别为6 0 0伏特和8 0 0伏特）Flyback in hard switching modewith 600V MOSFET使用6 0 0伏特场效应晶体管的硬开关反激变换器The hard switching approach(as shown in Fig. 26) doesn ctonsiderthe minimum drain-sourcevoltage. The MOSFET will be t urned on hard, in this caseat a voltage level of 500 V (at time point 3.3 以 s)The dischargeof circuits parasitic capacitanceseads to a high current spike during turning on.硬开关（图2 6所示）几乎不考虑漏源极电压的最小值。 场效应晶体管开通应力大，在这个例子里，开通电压在5 0 。伏特（在3 .3微秒的时间点）。由寄生电容引起的泄放电 流在开通时产生很高的电流尖刺。Fig. 26 Drain-sourcevoltage and drain current of hard switching 600V flyback图2 66 0 0伏特硬开关反激变换器的漏源极电压和漏极电流Flyback in quasi resonantmodewith 600 V MOSFET使用6 0 0伏特场效应晶体管的准谐振反激变换器The drain-sourcevoltage (Fig. 27) starts oscillating at the end of the flyback phaseand reachingthe minimum of 300 V when the MOSFET turns on.漏源极电压(图2 7)在反射过程结束后并减小到3 0 0 伏特时场效应晶体管导通。The duty cycle is lower comparedto an 800 V solution due to a lower reflectedvoltage of 100V. Shorter duty cycle for the sameoutput power results in higher peak currents on the primar y side.因为1 0。伏特的反射电压，比较8 0 0伏特解决方案 它有更小的占空比。小占空比实现同样的功率输生必须使用 更高的一次侧峰值电流。0E+箕II 后4 ES.E-OG G.E-OG 7.E-M： E-Fig. 27 Drain-sourcevoltage and drain current of quasi resonant600V flyback图2 76 0 0伏特准谐振反激变换器的漏源极电压和漏极电流Flyback in quasi resonantmodewith 800 V MOSFET使用8 0 0伏特场效应晶体管的准谐振反激变换器The drain-sourcevoltage (Fig. 28) starts oscillating at the end of the flyback phaseand reachingthe minimum of 100V when t he MOSFET turns on. The turning on current spike is low.漏源极电压(图2 8)在反射过程结束后并减小到1 0 0 伏特时场效应晶体管导通。开通电流尖刺比较低。The duty cycle is higher comparedo a 600V solution due t o a higher reflectedvoltage of 390V. Longer duty cycle for the s ame output power results in lower peak currents on the primary s ide.因为有3 9。伏特的反射电压，所以有比6 0 0伏特解 决方案更大的占空比。更大的占空比实现同样的输生功率可 以使用更低的一次侧峰值电流。 E+UOIE电62E-D63E-064 E-fle E-.E Q& 5.E4J6/E068 E OftFig. 28 Drain-sourcevoltage and drain current of quasi resonant800V flyback图2 88 0 0伏特准谐振反激变换器的漏源极电压和漏极电流Comparisoof spectra干扰频谱比较The spectra of the drain-sourcevoltagesfor correspondinglyba ck design(Fig. 26Fig. 27 and Fig. 28) are shown in Fig. 29.相应设计的反激变换器(图2 6、图2 7和图2 8)的漏源极电压干扰频谱如图2 9所示。牛图肠勤*卯出势曾案将务V T 。廿也缪 佛缈飞T4惚郭沙。088梆P且单疑专密上激事。中 革郭毛孑科左铮基/学一2审1P紫铮量/V*4 佛错革球y警热或器不。o8 湃中川舞触乎。夕唯病m OninbdjLf syBiy ui tun中拯 间叼 ui syns办 yiyai (ffuiyo 7f uvuosn tsimb 0，mp rnnmiutm avyoa ui s2nooo uo mn 叫 7 jrnoxs 伽m OmmbdMf ui sdjmydiuv soiuouuvy 啊她 o ，spi可 09 Jb 修，uy晒 n，叫加 加0g W / Buixun.注 痴SOK 叫7 uunp aPvyoa Bindiuvjo 在例如 nuofa(wai affpyoa a“nos-u卬即 须0g 眇 fo au也 xofmu 0g q pmiwjdbcd ag uvo 叼l，a如叩 /W09 ywg 07 yoAvdiuoo 27/ 3m90 u叫 Su 中28 9t jmv 27/J l 叫叫 sdioiidnbdjf 物 /叫网 n、ovq秘 uvuo sai xsvnb 何08 眇 fo uiru例ds dSvyoti 叩 usd? 四 us s/（淳的）静算曲受律密的 6 Z第(pdvjnuns) oBvyoa %unos-uya叩 眇 fo vjodds 6Z ,&Jb9*30 0013000110bHh9*30019*301010010001第二,0伏特，高于6 0 0伏特。它产生低频段的高振幅。开通发生在准谐振开关电压的最小值，这导致更高频段频谱 中更低的幅值。Due to the fact, that the 800V quasi resonantflyback has l ower peak current, its spectrumis significantly lower acrossalmos t completefrequencyrange.事实上，800伏特准谐振反激变换器拥有更低的峰值 电流，它的频谱意味着在全频带有更低的幅值。The 800V quasi resonantdesign with lower current peak and lower drain-sourcevoltage during turning on of the MOSFET de monstratesadvantagesin conductedEMI spectraregardingthe pri mary side.拥有更低峰值电流和场效应晶体管漏源极开通电压的8 0。伏特准谐振设计展示由一次侧传导电磁干扰降低的优 势。